AD8391AR(Rev0) View Datasheet(PDF) - Analog Devices

Part Name

Description

Manufacturer

AD8391AR
(Rev.:Rev0)

xDSL Line Driver 3 V to 12 V with Power-Down

Analog Devices 

AD8391

GENERAL INFORMATION

Theory of Operation

The AD8391 is a dual current feedback amplifier with high

output current capability. It is fabricated on Analog Devices’

proprietary eXtra Fast Complementary Bipolar Process (XFCB) that

enables the construction of PNP and NPN transistors with fT’s

greater than 3 GHz. The process uses dielectrically isolated

transistors to eliminate the parasitic and latch-up problems caused

by junction isolation. These features enable the construction of

high-frequency, low-distortion amplifiers.

The AD8391 has a unique pin out. The two noninverting inputs

of the amplifier are connected to the VMID pin, which is internally

biased by two 5 kΩ resistors forming a voltage divider between

+VS and –VS. VMID is accessible through Pin 7. There is also a

10 pF internal capacitor from VMID to –VS. The two inverting pins

are available at Pin 1 and Pin 8, allowing the gain of the amplifiers to

be set with external resistors. See the front page for a connection

diagram of the AD8391.

A simplified schematic of an amplifier is shown in Figure 3. Emitter

followers buffer the positive input, VP, to provide low-input current

and current noise. The low-impedance current feedback summing

junction is at the negative input, VN. The output stage is another

high-gain amplifier used as an integrator to provide frequency com-

pensation. The complementary common-emitter output provides the

extended output swing.

A current feedback amplifier’s bandwidth and distortion

performance are relatively insensitive to its closed-loop signal gain,

which is a distinct advantage over a voltage-feedback architecture.

Figure 4 shows a simplified model of a current feedback amplifier.

The feedback signal is an error current that flows into the inverting

node. RIN is inversely proportional to the transconductance of

the amplifier’s input stage, gmi. Circuit analysis of the pictured

follower with gain circuit yields:

VOUT

VIN

=

G × Tz(s)

Tz(s) + RF + G × RIN

where:

G = 1 + RF

RG

Tz(s) = RF

1 + sCT (RT )

RIN

=

1

gmi

≅ 125 Ω

Recognizing that G × RIN << RF , and that the –3 dB point is set

when Tz(s) = RF, one can see that the amplifier’s bandwidth

depends primarily on the feedback resistor. There is a value of

RF below which the amplifier will be unstable, as the amplifier

will have additional poles that will contribute excess phase shift.

The optimum value for RF depends on the gain and the amount

of peaking tolerable in the application. For more information

about current feedback amplifiers, see ADI’s High-Speed Design

Techniques at www.analog.com/technology/amplifiersLinear/

designTools/evaluationBoards/pdf/1.pdf.

VP

VN

VO

BIAS

Figure 3. Simplified Schematic

G=1

+

VO

VIN

RIN IIN

IT = IIN CT RT

+

–

VOUT

–

RF

RG

Figure 4. Model of Current Feedback Amplifier

Feedback Resistor Selection

In current feedback amplifiers, selection of the feedback and

gain resistors will impact distortion, bandwidth, noise, and gain

flatness. Care should be exercised in the selection of these resistors

so that the optimum performance is achieved. Table I shows the

recommended resistor values for use in a variety of gain settings for

the test circuits in TPC 1 and TPC 19. These values are only

intended to be a starting point when designing for any application.

Table I. Resistor Selection Guide

Gain

–1

–2

–3

–4

–5

RF (⍀)

909

909

909

909

909

RG (⍀)

909

453

303

227

178

–10–

REV. 0

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