APU3039 View Datasheet(PDF) - Advanced Power Electronics Corp

Part Name

Description

Manufacturer

APU3039

SYNCHRONOUS PWM CONTROLLER WITH OVER CURRENT PROTECTION

Advanced Power Electronics Corp 

Advanced Power

Electronics Corp.

APU3039

Choose AP9408AGH for control MOSFET and AP9412AGH These values are taken under a certain condition test.

for synchronous MOSFET. These devices provide For more details please refer to the AP9408AGH and

low on-resistance in a compact TO-252 package.

AP9412AGH data sheets.

The MOSFETs have the following data:

AP9408AGH

VDSS = 30V

ID = 53A

RDS(ON) = 10mΩ

AP9412AGH

VDSS = 30V

ID = 68A

RDS(ON) = 6mΩ

The total conduction losses will be:

P = P + P CON(TOTAL)

CON(UPPER)

CON(LOWER)

PCON(TOTAL) = 0.64W

The switching loss is more difficult to calculate, even

though the switching transition is well understood. The

reason is the effect of the parasitic components and

switching times during the switching procedures such

as turn-on / turnoff delays and rise and fall times. The

control MOSFET contributes to the majority of the switch-

ing losses in synchronous Buck converter. The synchro-

nous MOSFET turns on under zero voltage conditions,

therefore, the turn on losses for synchronous MOSFET

can be neglected. With a linear approximation, the total

switching loss can be expressed as:

t t PSW =

VDS(OFF)

2

x

r+

T

f x ILOAD

---(12)

Where:

VDS(OFF) = Drain to Source Voltage at off time

tr = Rise Time

tf = Fall Time

T = Switching Period

ILOAD = Load Current

The switching time waveform is shown in Figure 10.

VDS

90%

10%

VGS

td(ON)

tr td(OFF)

tf

Figure 10 - Switching time waveforms.

By using equation (12), we can calculate the total switch-

ing losses.

PSW(TOTAL) = 150mW

Programming the Over-Current Limit

The over-current threshold can be set by connecting a

resistor (RSET) from drain of low side MOSFET to the

OCSet pin. The resistor can be calculated by using equa-

tion (2).

The RDS(ON) has a positive temperature coefficient and it

should be considered for the worse case operation.

RDS(ON) = 8mΩx 1.5 = 12mΩ

ISET ≅ IO(LIM) = 8A x 1.5 = 12A

(50% over nominal output current)

This results to:

RSET = 5.76KΩ

Feedback Compensation

The APU3039 is a voltage mode controller; the control

loop is a single voltage feedback path including error

amplifier and error comparator. To achieve fast transient

response and accurate output regulation, a compensa-

tion circuit is necessary. The goal of the compensation

network is to provide a closed loop transfer function with

the highest 0dB crossing frequency and adequate phase

margin (greater than 45。).

The output LC filter introduces a double pole, –40dB/

decade gain slope above its corner resonant frequency,

and a total phase lag of 180。 (see Figure 11). The Reso-

nant frequency of the LC filter is expressed as follows:

1

FLC =

2π x LO x CO

---(13)

Figure 11 shows gain and phase of the LC filter. Since

we already have 180。 phase shift just from the output

filter, the system risks being unstable.

Gain

Phase

0dB

0。

-40dB/decade

From AP9408AGH data sheet we obtain:

AP9408AGH

tr = 5ns

tf = 6ns

-180。

FLC Frequency

FLC Frequency

Figure 11 - Gain and phase of LC filter.

10

Share Link: