MC13175D View Datasheet(PDF) - Motorola => Freescale

Part Name

Description

Manufacturer

MC13175D

UHF FM/AM TRANSMITTER

Motorola => Freescale 

MC13175 MC13176

Figure 16 shows the improved hold–in range of the loop.

The ∆fref is moved 950 kHz with over 200 µA swing of control

current for an improved hold–in range of ±15.2 MHz or

± 95.46 Mrad/sec.

Figure 16. MC13176 Reference Oscillator

Frequency versus Oscillator Control Current

10.6

Closed Loop Response:

10.4

fo = 32 x fref

VCC = 3.0 Vdc

ICC = 38 mA

10.2

Pout = 4.8 dB

Imod = 2.0 mA

10

Vref = 500 mVp–p

9.8

9.6

9.4

–150

–100

– 50

0

50

100

I6, OSCILLATOR CONTROL CURRENT (µA)

Lock–in Range/Capture Range

If a signal is applied to the loop not equal to free running

frequency, ff, then the loop will capture or lock–in the

signal by making fs = fo (i.e. if the initial frequency

difference is not too great). The lock–in range can be

expressed as ∆ωL ~ ± 2∂ωn

FM Modulation

Noise external to the loop (phase detector input) is

minimized by narrowing the bandwidth. This noise is minimal

in a PLL system since the reference frequency is usually

derived from a crystal oscillator. FM can be achieved by

applying a modulation current superimposed on the control

current of the CCO. The loop bandwidth must be narrow

enough to prevent the loop from responding to the

modulation frequency components, thus, allowing the CCO

to deviate in frequency. The loop bandwidth is related to the

natural frequency ωn. In the lag–lead design example where

the natural frequency, ωn = 5.0 krad/sec and a damping

factor, ∂ = 0.707, the loop bandwidth = 1.64 kHz.

Characterization data of the closed loop responses for both

the MC13175 and MC13176 at 320 MHz (Figures 7 and 8,

respectively) show satisfactory performance using only a

simple low–pass loop filter network. The loop filter response

is strongly influenced by the high output impedance of the

push–pull current output of the phase detector.

fc = 0.159/RC;

For R = 1.0 k + R7 (R7 = 53 k) and C = 390 pF

fc = 7.55 kHz or ωc = 47 krad/sec

The application example in Figure 18 of a 320 MHz FM

transmitter demonstrates the FM capabilities of the IC. A high

value series resistor (100 k) to Pin 6 sets up the current

source to drive the modulation section of the chip. Its value is

dependent on the peak to peak level of the encoding data

and the maximum desired frequency deviation. The data

input is AC coupled with a large coupling capacitor which is

selected for the modulating frequency. The component

placements on the circuit side and ground side of the PC

board are shown in Figures NO TAG and NO TAG,

respectively. Figure 20 illustrates the input data of a 10 kHz

modulating signal at 1.6 Vp–p. Figures 21 and 22 depict the

deviation and resulting modulation spectrum showing the

carrier null at – 40 dBc. Figure 23 shows the unmodulated

carrier power output at 3.5 dBm for VCC = 3.0 Vdc.

For voice applications using a dynamic or an electret

microphone, an op amp is used to amplify the microphone’s

low level output. The microphone amplifier circuit is shown in

Figure 17. Figure 19 shows an application example for NBFM

audio or direct FSK in which the reference crystal oscillator is

modulated.

Figure 17. Microphone Amplifier

Data

VCC

Input

100k 120k

3.3k 3.9k 1.0

VCC

Voice

Input

Electret

Microphone

1.0k 10k

10k

MC33171

Data or

Audio

Output

Local Oscillator Application

To reduce internal loop noise, a relatively wide loop

bandwidth is needed so that the loop tracks out or cancels

the noise. This is emphasized to reduce inherent CCO and

divider noise or noise produced by mechanical shock and

environmental vibrations. In a local oscillator application the

CCO and divider noise should be reduced by proper

selection of the natural frequency of the loop. Additional low

pass filtering of the output will likely be necessary to reduce

the crystal sideband spurs to a minimal level.

10

MOTOROLA RF/IF DEVICE DATA

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